Decoder

ABSTRACT

An H-ARQ system wherein the transmission of two consecutive, or sequential, blocks of information bits are considered jointly; i.e., one of the blocks of information being embedded within the other one of the blocks of information. If a retransmission for the first block is necessary, the system processes both blocks jointly. The system is provided with cross-packet coding which extends current H-ARQ schemes for point-to-point communications wherein the transmission of two consecutive block of information bits is considered jointly. If a retransmission for the first block is necessary, the system processes both blocks jointly. This allows both blocks to be decoded without errors at the receiver after the retransmission.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part application of, and claims priority from and incorporates by reference, U.S. Non-Provisional application Ser. No. 11/699,792 filed on Jan. 30, 2007, entitled, “Embedded Retransmission Scheme with Cross-Packet Coding,” which claimed priority from and incorporated by reference U.S. Provisional Application Ser. No. 60/796,514, filed May 1, 2006, entitled, “Embedded Retransmission Scheme with Cross-Packet Coding.” This application also claims priority from, and incorporates by reference, U.S. Provisional Application Ser. No. 60/888,658, filed Feb. 7, 2007.

TECHNICAL FIELD

This invention relates generally to data transmission systems and more particularly to data transmission systems wherein data may be required to be re-transmitted.

BACKGROUND

In wireless communication, channel fading and interference noise fluctuate rapidly according to the channel conditions. It is well known that the packet errors occur when the attempted transmission rate is higher than the acceptable channel rate. Several techniques have been used to alleviate this problem. Link adaptation adjusts the transmission rate (and the amount of redundancy) to compensate for this fluctuation. However, it is assumed that the channel is stationary during the adaptation and transmission periods. Whenever error occurs, packet retransmission is often used to ensure reliable packet delivery. In one type of system it may be required to retransmit data. One such system is a Hybrid ARQ techniques such as incremental redundancy and Chase combining have been proposed to improve the spectral efficiency of retransmission. For delay-sensitive applications such as voice and video, retransmission also causes additional delay, which might impact the quality. Thus, in a typical wireless communication system, a packet retransmission is used to recover from channel errors at the cost of additional delay and overhead (spectral efficiency). In addition, for delay-sensitive applications the arrival of retransmitted packets may be too late to be used at the codecs.

More particularly, in current H-ARQ schemes, a channel encoder encodes a block of information bits u₁, and outputs a block of code bits x₁ which is sometimes referred to as a codeword. The first transmission x₁₁ to the receiver contains a part of the codeword. If the receiver cannot decode the codeword without error (the error detection can be done with a cyclic redundancy check (CRC)), a retransmission from the transmitter is necessary. While in ARQ schemes of type I (separate ARQ and FEC), the transmission starts again from the beginning, in H-ARQ schemes the receiver stores the first transmission and the transmitter sends another part of the codeword x₁₂ in the second transmission. Then, both parts of the codeword which were received are used for decoding.

In previous works (see J. Nonnenmacher, E. Biersack, and D. Towsley, “Parity-based loss recovery for reliable multicast transmission”, ACM SIGCOMM Computer Communication Review, vol. 27, pp. 289-300, October 1997 and H. Lundqvist and G. Karlsson, “TCP with End-to-End Forward Error Correction” Technical Report, TRITA-IMIT-LCN R 03:03, ISSN 1651-7717, ISRN KTH/IMIT/LCN/R-03103-SE, KTH, Royal Institute of Technology, Sweden) with automatic repeat request (ARQ) and forward error correction (FEC) on the packet level, it is assumed that FEC on the bit level delivers an erasure channel for the packet level. On the packet level a second FEC is done with the complete packets.

FIG. 1 shows a conventional H-ARQ system for transmitting two blocks of information bits u₁ and u₂. The transmitter wants to transmit information bits which are segmented into blocks of K bits to the receiver. A channel encoder processes a first block of information bits u₁ and outputs a block of N code bits x₁ which we call codeword. The first transmission x₁₁ to the receiver contains a punctured version of the codeword which consists of M code bits. The puncturing is done by a H-ARQ functionality. At the receiver, y₁₁, which is a disturbed version of x₁₁, is received. The H-ARQ functionality at the receiver transforms y₁₁ to y₁₁. The parts in y₁ which correspond to bits of x₁ which are not transmitted are filled with the log-likelihood value of 0 because there is no information available at the receiver about these bits. If the receiver cannot decode the codeword without error (an integrity check can be done with Cyclic Redundancy Check), a retransmission x₁₂ from the transmitter is necessary. Note that the error detection can be done with a cyclic redundancy check (CRC) which has to be attached to the block u₁ before encoding. The receiver stores the first transmission y₁₁ and the transmitter sends another subset x₁₂ with M code bits in the second transmission. Both parts of the codeword which were received are used for decoding like it is described in a paper by J. Hagenauer. Rate-Compatible Punctured Convolutional Codes (RCPC Codes) and their Applications. IEEE Trans. on Communications, 36(4):389-400, April 1988. This H-ARQ scheme is called incremental redundancy because the amount of redundancy increases with the retransmission. The next block of information bits, u₂ is treated separately from the first block u₁. The process is summarized in the flow diagram of FIG. 1A.

SUMMARY

In accordance with the present invention, a method is provided for transmitting information comprising transmitting a first block of information; and transmitting the first block of information jointly (i.e., embedded) with a second block of information.

In one embodiment, a method is provided for transmitting information comprising: transmitting a first block of information; and transmitting the first block of information, the first block of information being embedded with a second block of information.

In one embodiment, a method is provided for transmitting information comprising: transmitting a first block of information; and transmitting the first block of information, bits of the first block of information being embedded into bits of a second block of information.

In one embodiment, a method is provided for transmitting information comprising: transmitting a first block of information; and transmitting the first block of information, bits of the first block of information being combined with bits of a second block of information.

In one embodiment, a method is provided for transmitting information comprising: transmitting a first block of information; and transmitting the first block of information, with each one of the bits of the first block of information being combined with a corresponding one of bits of a second block of information.

In one embodiment, the invention, a method is provided for transmitting information comprising: transmitting a first block of information; detecting errors in the first block of information; and, if errors are detected, transmitting the first block of information embedded within a second block of information.

In one embodiment, a method is provided for transmitting information in a H-ARQ system comprising: embedding re-transmission of a first block of information with a second block of information if retransmission for the first block is necessary.

In one embodiment, the embedded retransmission of the first block of information is combined using XOR or soft combining with probabilistic approach.

With such method additional transmission delay or transmission bandwidth is eliminated while maintaining the capability of re-transmitting the erroneous packets.

In one embodiment, if a retransmission is required for the first block of information, the process transmits both the first block of information and a second block of information embedded with the first block of information.

In one embodiment, an H-ARQ system is provided wherein the transmission of two consecutive, or sequential blocks of information bits are considered jointly; i.e., embedded one block with the other. If a retransmission for the first block is necessary, the system processes both blocks jointly; i.e., the first block embedded within the second block.

With such an arrangement, a H-ARQ system is provided with cross-packet coding which extends current H-ARQ schemes for point-to-point communications wherein the transmission of two consecutive, or sequential block of information bits is considered jointly; i.e., embedded one block with the other. If a retransmission for the first block is necessary, the system processes both blocks jointly; i.e., the first block embedded within the second block. This allows both blocks to be decoded without errors at the receiver after the retransmission.

In one embodiment, an encoder and a decoder are provided to decode iteratively for a system with H-ARQ with cross-packet coding.

With such H-ARQ with cross-packet coding, the second transmission x₁₂ which is normally only used for the decoding of the first block of information bits u₁ is processed jointly with a new block of information bits u₂ and thus allowing both to support the transmission of u₁ and the transmission of u₂ jointly; i.e., the transmission of u₁ embedded within the transmission of u₂.

In one embodiment, FEC is done still on the bit level even if the coding for two 20 packets is considered jointly.

In one embodiment, the H-ARQ with cross-packet coding the second transmission x₁₂ which is normally only used for the decoding of the first block of information bits u₁ is processed jointly with a new block of information bits u₂ and thus, allow both to support the transmission of u₁ and the transmission of u₂.

In one embodiment, a method is provided for transmitting information in a H-ARQ system comprising: embedding re-transmission of a first block of information with a second block of information if retransmission for the first block is necessary where codewords for both the first transmission and the embedded re-transmission are decoded iteratively at the receiver.

In one embodiment, an encoder and a decoder are provided which can be decoded iteratively for a system with H-ARQ with cross-packet coding.

In one embodiment, the embedded retransmission of the erroneous packet is combined with the new packet to be transmitted. The embedment can be done with hard combining using XOR or soft combining with probabilistic approach. The point of packet combining can be done at various points including at the input of the channel encoder, at the output of the channel encoder, or at the output of the channel modulator. Here the channel modulator block is omitted. The channel modulator is the last block in the transmitter chain before the signal is sent out over the medium.

When the receiver cannot successfully decode the packet, it automatically triggers the retransmission. At the transmitter, it sends new information with embedded retransmitted information. Since the receiver knows a priori information, it can extract some of embedded information from the newly received packet and start the decoding process or use a priori information to assist decoding. Some of the residue at this stage can be considered as interference. If the decoding of new information is successful, the receiver can then use this information to decode retransmitted information. Chase combining may also be used to combine the original transmission and the re-transmission.

The details of one or more embodiments of the invention are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the 20 invention will be apparent from the description and drawings, and from the claims.

DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram of a Hybrid Automatic Repeat-reQuest (H-ARQ) system according to the PRIOR ART;

FIG. 1A is a flow diagram of the process used by the Hybrid Automatic Repeat-reQuest (H-ARQ) system of FIG. 1 according to the PRIOR ART;

FIG. 2 is a block diagram of a Hybrid Automatic Repeat-reQuest (H-ARQ) system according to the invention;

FIG. 2A is a flow diagram of the process used by the Hybrid Automatic Repeat-reQuest (H-ARQ) system of FIG. 1 according to the invention;

FIG. 3 is a block diagram of a cross packet channel encoder used in the system of FIG. 2 wherein a point of packet combining is performed at the input of the channel encoder;

FIG. 4 is a block diagram of a cross packet channel decoder used in the system of FIG. 2;

FIGS. 5A-5G are diagrams useful in understanding redundancy-free retransmission used in the Hybrid Automatic Repeat-reQuest (H-ARQ) system according to the invention;

FIG. 6 is a block diagram of a transmitter of a Hybrid Automatic Repeat-reQuest (H-ARQ) system according to the invention wherein a point of packet combining is performed at the input of the channel encoder;

FIG. 7 is a block diagram of a receiver with the transmitter of FIG. 6 in a Hybrid Automatic Repeat-reQuest (H-ARQ) system according to the invention.

FIG. 8 is a block diagram of a cross packet channel encoder according to another example embodiment.

FIG. 9 is a block diagram of a cross packet channel encoder according to another example embodiment.

FIG. 10 is a block diagram of a cross packet channel decoder according to another example embodiment.

FIG. 11 is a block diagram of a cross packet channel decoder according to another example embodiment.

FIG. 12 is a block diagram of an apparatus according to an example embodiment.

FIG. 13 is a flowchart showing a method according to an example embodiment.

FIG. 14 is a flowchart showing a method according to another example embodiment.

Like reference symbols in the various drawings indicate like elements.

DETAILED DESCRIPTION

Referring now to FIG. 2, a transmission system 10 for transmitting two blocks of information bits u₁ and u₂ is shown with H-ARQ and cross-packet coding. Briefly, the first transmission x₁₁ is the same as in the system without cross-packet coding (FIG. 1). Thus, the transmitter 12 wants to transmit information bits u₁ which are segmented into blocks of K bits to a receiver 14. A channel encoder 16 processes the first block of information bits u₁ and outputs a block of N code bits x₁ which we call codeword. The first transmission x₁₁ to the receiver 14 contains a punctured version of the codeword which consists of M code bits. The puncturing is done by a H-ARQ 18 functionality. At the receiver 14, y₁₁, which is a disturbed version of x₁₁, is received. The H-ARQ functionality 20 at the receiver 14 transforms y₁₁ to y₁. The parts in y₁ which correspond to bits of x₁ which are not transmitted are filled with the log-likelihood value of 0 because there is no information available at the receiver about these bits. If the receiver 14 cannot decode the codeword without error (an integrity check can be done with Cyclic Redundancy Check) with the channel decoder 40, a retransmission x₁₂ from the transmitter 10 is necessary. Note that the error detection can be done with a cyclic redundancy check (CRC) which has to be attached to the block u₁ before encoding. Here, however, if the receiver 14 requires a retransmission, u₁ is allowed to be retransmitted jointly, i.e., embedded within, the initial transmission of the second block of information bits to be transmitted, u₂ using cross-packet coding. There are a number of possible realizations of this joint transmission such as those shown in FIGS. 3, 5 and 6, to be described. The cross-packet channel encoder 22, shown and to be described in more detail in FIG. 3, outputs the code word x₃ which is based on the both inputs u₁ and u₂. The H-ARQ 24 functionality x₃ and outputs the bits are sent in the second transmission to the receiver. The H-ARQ 26 functionality at the receiver 14 transforms the channel output into y₃. The cross-packet channel decoder 28, shown and to be described in more detail in FIG. 4, delivers estimates û₁ and û₂ about both blocks of information bits based on y₁ and y₃. It is also possible to provide a third transmission. Again, here both blocks u₁ and u₂ consists of the same number of information bits K and that all transmissions x₁₁, x₃₁ and x₃₂ contain the same number of code bits M. Thus, when the receiver 14 cannot successfully decode the packet, it automatically triggers the retransmission. The transmitter 10 sends new information u₂ with embedded retransmitted information u₁. Since the receiver 14 knows a priori information about u₁ from the previous transmission, it can extract some embedded information from the newly received packet and start the decoding process or use a priori information to assist decoding. Some of the residue at this stage can be considered as interference and may impact the decoding performance of u₂. After the first decoding, the receiver 14 gets both estimates u₁ and U 2 from the channel decoder 42. The new estimate u₁ can then be used as a priori information for the channel decoder 40. It is possible to iteratively exchange information about u₁ and u₂ between the two channel decoders 40 and 42 to improve the successful decoding probabilities of u₁ and u₂. Chase combining may also be used to combine the original transmission and the re-transmission. The process is summarized in the flow diagram of FIG. 2A.

Cross-Packet Channel Encoder 22

Referring now to FIG. 3, the cross-packet channel encoder 22 is shown. Both blocks of information bits u₁ and u₂ are inter-leaved in interleavers 30, and 302, respectively, and appear then alternatively (e.g., time multiplexed) in multiplexer 32) at the input of a channel encoder 34. The input of the channel encoder 34 is termed u₃. The input u₃ contains 2 K bits. The block u₃ is channel encoded to obtain the output x₃. The output x₃ of the cross-packet channel encoder 22 has approximately double length compared to the output of a conventional channel encoder 16. However, more bits can be punctured in the H-ARQ block which outputs x₃₁ with the same size as x₁ (M bits). To reduce the complexity at the H-ARQ block 24, the channel encoders 16, 22 are systematic. However, the channel encoders 16 and 22 are not required to be systematic.

Cross-Packet Channel Decoder 28

Referring now to FIG. 4, the cross-packet channel decoder 28 is shown. Here, the cross-packet channel decoder 28 applies the turbo principle in a paper by J. Hagenauer. The Turbo principle: Tutorial introduction and state of the art. In Proc. International Symp. on Turbo Codes and Related Topics, Brest, France, September 1997. The cross-packet channel decoder 28 delivers the estimates u₁ and U 2 and based on the channel outputs y₁ and y₃.

First, the channel decoder 40 which corresponds to the turbo encoder of u₁ calculates extrinsic information i.e. log likelihood ratios L_(e)(u₁) from its input y₁. Then, the second channel decoder 42 which corresponds to the encoder of u₃ tries to decode u₃ based on the channel output y₃ and on a priori information about u₁ which is provided from the first channel decoder 40. The extrinsic information L_(e)(u₁) has to be interleaved in interleaver 36 and mixed with zeros in mixer 44 in order to obtain a priori information for u₃. After a de-interleaver 39, the second channel decoder 42 can deliver a priori information L_(a) (u₁) about u₁ which can be exploited again by the first channel decoder 40. It is possible to apply the turbo principle and to iteratively exchange soft information about u₁ between the two decoders 40, 42 several times.

The decoders 40, 42 are here soft-in/soft-out decoders such as those used in the convolutional codes in the parallel concatenated convolutional code (PCCC) channel encoder which uses a linear approximation as in a paper by J.-F. Cheng and T. Ottosson. Linearly approximated log-map algorithms for turbo coding. In Proc. IEEE Vehicular Technology Conference (VTC), Houston, Tex., May 2000 to the optimal log-MAP algorithm in decoders 40 and 42. This approximation is compared with other versions of the log-MAP algorithm in the paper by M. C. Valenti and J. Sun. The Universal Mobile Telecommunications System (UMTS) Turbo Code and an Efficient Decoder Implementation Suitable for Software-Defined Radio. Intern. Journal of Wireless Information Networks, 8(4):203-215, October 2001. According to the paper by M. C. Valenti and J. Sun, the linear approximation provides a good trade-off between accuracy and complexity.

Example of channel code used in cross packet coding (Convolutional Turbo Code) As noted above, the cross-packet channel encoder 22 and decoder 28 are shown in FIGS. 3 and 4, respectively. Here, a specific code design is used for the cross-packet channel encoder 22 and decoder 28. The code design is related to the turbo network code which is described in a paper by C. Hausl and J. Hagenauer, Iterative network and channel decoding for the two-way relay channel. In Proc. IEEE International Conference on Communications (ICC 06), Istanbul, Turkey, June 2006. Whereas in the paper a wireless communication network with three nodes is considered and two blocks of data which were received from other nodes are encoded jointly, i.e., one block embedded one within the other block, here a point-to-point communication is used where two blocks from the same node are encoded jointly. The code design can be still useful to gain diversity with cross-packet coding. The proposed code can be decoded efficiently in an iterative way.

Iterative Cross-Packet Channel Decoding

Here, the method in this example uses a specific code design for the cross-packet channel encoder and decoder. The code design is related to the turbo network code which is described in C. Hausl and J. Hagenauer. Iterative network and channel decoding for the two-way relay channel. In Proc. IEEE International Conference on Communications (ICC 06), Istanbul, Turkey, June 2006.

Cross-Packet Channel Encoder

Referring to both FIG. 2 and FIG. 3, here, in this example both the channel encoder 16 which encodes u₁ and the channel encoder 34 in the cross-packet channel encoder 22 which encodes u₃ is chosen to be the parallel concatenated convolutional code (PCCC) which is used in UMTS (see 3GPP TR 25.944. Channel Coding and Multiplexing Examples (Release4), June 2001. This code including the required interleaver embedded in PCCC is described in M. C. Valenti and J. Sun. The UMTS Turbo Code and an Efficient Decoder Implementation Suitable for Software-Defined Radio. Intern. Journal of Wireless Information Networks, 8(4):203-215, October 2001. The additional interleaver 30 ₁ in FIG. 3 is determined according to M. C. Valenti and J. Sun. The UMTS Turbo Code and an Efficient Decoder Implementation Suitable for Software-Defined Radio. Intern. Journal of Wireless Information Networks, 8(4):203-215, October 2001. It is noted that the interleaver 30 ₂ is not used in this example.

Referring to again to FIG. 3, i.e., the cross-packet channel encoder 22, the information bits of an interleaved version of packet u₁ and the bits in the packet u₂ appear alternately at the input of a channel encoder 34. The input of the channel encoder 34 is termed u₃. The packet u₃ is channel encoded to obtain the output x₃.

The H-ARQ functionality after both the channel encoder 16 and the cross-packet channel encoder 22 uses regular puncturing schemes, similar to the one described in D. N. Rowitch and L. B. Milstein. On the Performance of Hybrid FECIARQ Systems Using Rate Compatible Punctured Turbo (RCPT) Codes. IEEE Trans. on Communications, 48(6):948-959, June 2000, to choose for each transmission M out of all code bits. The only differences after cross-packet encoding are that we always puncture the systematic bits of x₃ that correspond to u₁ because they are already included in the first transmission x₁₁ and that stronger puncturing is necessary.

Cross-Packet Channel Decoder 28

Referring again to FIG. 4, the cross-packet channel decoder 28 delivers the estimates u₁ and U 2, based on the channel outputs y₁ and y₃. First, the channel decoder 40 corresponding to the turbo encoder 16 of u₁ calculates extrinsic information i.e. log likelihood ratios L_(e)(u₁) from its input y₁. Then, the second channel decoder 42, which corresponds to the encoder 34 of u₃ tries to decode u₃ based on the channel output y₃ and on a priori information about u₁ provided from the first channel decoder 40. Once the decoder 42 has knowledge of u₁ and u₃, u₂ can be extracted. The extrinsic information L_(e)(u₁) has to be interleaved and mixed with zeros in order to obtain a priori information for u₃. The second channel decoder 42 can deliver a priori information L_(a)(u₁) about u₁ which can be exploited again by the first channel decoder 40. It is possible to apply the turbo principle and iteratively exchange soft information about u₁ between the two decoders 40,42 several times. Since the interleaver 30 ₂ in this example is not used, the de-interleaver 43 is not used as well.

The soft-in/soft-out decoders 40, 42 for the convolutional codes in the PCCC uses a linear approximation, see J.-F. Cheng and T. Ottosson. Linearly approximated log-map algorithms for turbo coding. In Proc. IEEE Vehicular Technology Conference (VTC OO), pages 2252-2256, Houston, Tex., May 2000, to the optimal log-MAP algorithm. All PCCC channel decoders 40 and 42 use ten iterations and the cross-packet channel decoder uses two iterations.

Extension to Other Channel Codes

It is now relatively straightforward to use other channel codes with this invention given that the code used in the encoder 22 and the decoder 28 allow an exchange of decoding information as previously described. Examples of such codes are Block Turbo Code and Low Density Parity Check Code (LDPC).

More particularly, y₁, is the received version of x₁₁. The H-ARQ 20 (FIG. 2) produces y₁, which is a reconstructed version of x₁. The channel decoder 50 (FIG. 2) then 10 tries to reconstruct u₁. If there is no error (an integrity check can be done with Cyclic Redundancy Check), the receiver 14 informs the transmitter 12 of successful reception. The transmitter 12 then sends new information and the decoder 50 performs the operation in the same way. If errors occur, the receiver 14 requests a retransmission from the transmitter 12 and keep the information relevant to y₁ (or the estimate of u₁) for further processing. The transmitter 12 then forwards both retransmitted and new information to the encoder 22 jointly. Interleavers 30 ₁ and 30 ₂ (FIG. 3) may be used to scramble information for better coding performance. Note that the scrambling sequences of interleavers 30 ₁ and 30 ₂ may be different. The channel encoder 22 generates another codeword x₃, which is typically larger than x₁. The H-ARQ 24 is then used again to reduce the size of this codeword to fit the allocated bandwidth and transmit x₃₁.

Further, y₃₁ is the received version of x₃₁ and H-ARQ 26 generates y₃ as the estimate of x₃. The channel decoder 28 uses a priori information of y₁ (or the estimate of u₁) to aid decoding along with y₃, estimates u₂, and generates the new estimate of u₁. The decoder 28 then again uses previously stored values of y₁, along with the new estimate of u₁ as a priori information, and tries to get the new estimate of u₁. The process can be repeated in an iterative manner.

Before discussing the system of FIG. 6, reference is made to FIGS. 5A-5G which illustrate an example of redundancy-free retransmission. In this example, hard combining of coded information is used with Chase combining. FIG. 5A shows the transmission of the original packet 10 and corresponding errors. The receiver stores the erroneous packet 100 and waits for retransmission. Next, the transmitter combines the previous transmission 10 with the current one 20 with a simple XOR operation (hard combining) (FIG. 5B). Next, the transmitter also informs the receiver of this operation and transmits the packet. The packet 200 may arrive at the receiver with some errors due to channel noise and interference. The receiver performs an XOR operation to extract new information as shown in FIG. 5C. After forward error correction, there might be some uncorrectable errors in the packet 300 and the receiver performs another XOR operation to obtain information about the retransmitted packet 10 as illustrated in FIG. 5D. Next, the receiver combines information from the first and second transmission and decodes the packet as shown in FIG. 5E. Chase combining may be used for this purpose. Assuming the decoding is successful, correct decoding of the retransmitted packet 10 is obtained. Similarly, in order to decode the packet 20, the receiver repeats steps in FIG. 5C with the correctly decoded packet 10 to get a new estimate of the packet 20 before combining a new estimate with the previously decoded version 300 as shown in FIGS. 5F and 5G. Assuming successful decoding, the packet 20 can be decoded without error. It is possible to iteratively repeat steps shown in FIGS. 5C-5G to improve decoding performance.

It should be noted that this example only illustrates the concept of redundancy-free retransmissions and this invention is not limited to the example given here.

Referring now to FIG. 6, an encoder at the transmitter 200 of a system with cross-packet coding (XOR) after channel encoding is shown. First and the second block of information bits u₁ and u₂, are passed to channel encoders 202 ₁ and 202 ₂, respectively, to obtain encoded first and second blocks of codewords x₁ and x₂, respectively. The codewords x₁ are fed to a Hybrid Automatic Repeat-reQuest (H-ARQ) system 204 ₁ for transmission to a receiver 400 (FIG. 7) as codeword x₁₁. If a retransmission is needed, the codewords x₁ are fed to XOR 206 along with the second codewords x₂ after such second codewords are passed through an interleaver 208, as shown, the output of the XOR 206 producing a new block x₃. The new block x₃ is fed to a Hybrid Automatic Repeat-reQuest (H-ARQ) system 204 ₂ for transmission to the receiver 400 (FIG. 7) as codeword x₃₁.

Thus, in FIG. 6, u₁ and u₂ first encoded into the outputs x₁ and x₂ and the outputs x₁ and x₂ are combined by the XOR 206 after channel encoding to obtain the output x₃. It is noted that there are different possibilities as to when to combine the first and the second block of information bits u₁ and u₂. For example, they can be combined by an XOR before channel encoding resulting in a new block u₃ with the block u₃ being channel encoded to obtain output x₃ and a part x₃₁ of x₃ is sent as the second transmission to the receiver 400 (FIG. 7). In both cases, the information is combined with XOR 206 (e.g. x₃=x₁⊕x₂). Also the blocks (e.g. x₃=x₁⊕Π(x₂)) can be interleaved or the blocks (e.g. x₃=Π([x₁x₂]) may be mixed. In general, all possible linear combinations can be denoted as x₃=G₁u₁⊕G₂u₂ where G₁ and G₂ are linear operators.

Thus, referring to FIG. 6, the blocks of information bits u₁ and u₂ are encoded in to obtain the outputs x₁ and x₂. The first transmission x₁₁ to receiver 400 (FIG. 7) is a part of x₁. If the receiver 400 requests a retransmission, the first codeword x₁ and an interleaved version of the second codeword Π(x₂) are combined to x₃=x₁⊕Π(x₂) so that the transmitter 200 transmits the part x₃₁ of x₃ to the receiver 400.

The receiver 400 (FIG. 7) obtains the channel output y₁₁ which is the disturbed version of x₁₁. The H-ARQ 402 ₁ functionality transforms y₁₁ to y₁. The parts of the codeword which are not in y₁₁ are filled with the soft value 0. The channel decoder 410 ₁ obtains the estimate û₁ from its input y₁. If û₁ is detected to be erroneous, the receiver 400 requests a retransmission from the transmitter 200 (FIG. 6). The receiver 400 then receives the channel output y₃₁ which is the disturbed version of x₃₁. The H-ARQ 4022 functionality transforms y₃₁ to y₃.

In order to obtain an estimate about the second block of information bits û₂, the process de-interleaves the output of a box-plus (BOX) operation 404 ₂ (such as that described in J. Hagenauer, E. Offer, and L. Papke, “Iterative Decoding of Binary Block and Convolutional Codes”, IEEE Trans. on Information Theory, vol. 42, no. 2, pp. 429-445, March 1996) of y₃ and L_(e)(x₁) and obtain a priori information L_(a)(x₂)=Π⁻¹(y₃BOXL_(e)(x₁)) about the codeword x₂ for the channel decoder 410 ₂. L_(e)(x₁) is extrinsic information about the codeword x₁ which is output from the first channel encoder 202 ₁ (FIG. 6).

It is noted that it is possible to iteratively exchange soft information about x₁ and x₂ between the two decoders 410 ₁ and 410 ₂. In such case, the input to decoder 410 ₁ uses a priori information L_(a)(x₁)=(y₃ΠBOXL_(e)(x₂)) about the codeword x₁ for the channel decoder 410 ₁. L_(e)(x₂) is extrinsic information about the codeword x₂ which is output from the second channel encoder 202 ₂ (FIG. 6). Then the two decoders 410 ₁ and 410 ₂ iteratively exchange information as described above. The values in y₃ will not change during the decoding process and the output of the box plus operation 404 ₁ and 404 ₂ is always limited by the worse of the two inputs.

In operation, when a packet containing data bits, which is transmitted over a medium (wired or wireless), is corrupted due to noise and/or interference, a retransmission is normally needed in order to maintain data's integrity, i.e., files needed to be perfectly reconstructed, the system allows the transmitter to multiplex retransmissions of a corrupted packet with the next transmission of new data, thus reducing the need of additional resources and end-to-end delay. Specifically channel coding and/or data multiplexer can be designed in a way that more than one source of data be combined to form a single packet (cross-packet coding). The receiver is able to extract different data sources with the knowledge of the initial and combined transmissions. It is possible to remove feedback altogether. The idea is to multiplex information with minimal impact on regular transmissions (non-retransmissions). If the first packet is received correctly, the receiver should be able to remove additional information with negligible penalty. Thus a feedback mechanism may not be necessary. Thus, a point-to-point system detection of an error to determine whether a retransmission is required is not necessarily and the first block may be embedded with the second block independent of a detected error. Further, the invention may be used in broadcast or multicast system without detection of an error and in, such case the first block would be embedded with the transmission of the second block.

In another embodiment, the same codeword x₁₁ is used for both initial transmission and retransmission. Then at the receiver, soft combining techniques such as Chase combining may be used to jointly decode both transmissions.

It is noted that the use of XOR(⊕) or probabilistic combining allows additional information to be multiplexed with regular transmissions. If the first transmission is too noisy, it is possible that the extrinsic information of x₁₁ is unreliable and L_(a)(x₂), a priori information from the previous encoding contains too much residue from the first transmission. This causes error propagation from one codeword to another as described earlier. However if the channel is fully interleaved and the link adaptation is working properly (preferably conservative), the packet errors tend to be bursty and do not span over multiple packets. Therefore the likelihood of successful joint decoding is high due to additional diversity gained from the second retransmission.

It is further noted that this invention can be extended to systems that allow more than one retransmission. Specifically, if the second decoding fails, the transmitter may try to embed the first and second messages to the third transmission. It should also be noted that the system requires additional overhead for acknowledgments. In scenarios where no acknowledgment is used such as the transmission of UDP or multicast packets, it is possible to use H-ARQ with cross-packet coding without any acknowledgement. The first transmission can always be transmitted together with the second transmission. If the first decoding is successful as indicated by the CRC, the residue (the first transmitted codeword) on the second transmission can be completely removed. If not the joint decoding can be used to increase the possibility of successful decoding.

FIG. 8 is a block diagram of a cross packet channel encoder 22 according to another example embodiment. In an example embodiment, one mother code may encode one information packet (e.g., u₁ as it passes through the channel encoder 16 shown in FIG. 2), while another mother code may encode two information packets (e.g., u₁ and u₂ as they pass through either channel encoder shown in FIG. 8 or FIG. 9). The two packets, such as u₁ and u₂, may be encoded using low density parity check coding (LDPC), according to an example embodiment. This encoding may generate the new mother codeword x₃.

According to an example embodiment, the two packets may be encoded using two steps. FIG. 9 is a block diagram of a cross packet channel encoder 22 according to another example embodiment. In this example, u₁ may be concatenated, u₂ may be concatenated, and the concatenated u₁ may be interleaved with the concatenated u₂; or, u₁ may be concatenated and interleaved with u₂. While FIG. 9 shows the interleaving by alternating the concatenated u₁ bits with the concatenated u₂ bits, other methods of concatenation may be used. The interleaving may, for example, remove the structure of practical LDPC codes, such as those found in Section 8.4.9.2.5 of the IEEE 802.16e WiMAX standard. The referenced codes may, for example, include block matrices where each non-zero block is a shifted identity matrix, which may allow for efficient LDPC encoding. In an example in which random LDPC codes are used, the LDPC matrix may be inherently random, and the interleaving may be skipped.

The information packets may, for example, be encoded using two convolutional codes applied to different permutations of the packet, such as Turbo coding, or any other linear code. Decoding these coded packets may make use of the iterative algorithm such as the Turbo principle, by passing decoded information between two soft-input/soft-output decoders (for the two constituent convolutional codes), according to an example embodiment.

The codewords x₁ and x₃ may be transmitted over a wired or wireless medium. The codewords x₁ and x₃ may be transmitted as fragments, according to an example embodiment; the “fragments” may include punctured versions of the codewords x₁ and x₃, or may include segments of the codewords x₁ and x₃ broken into smaller pieces and transmitted separately. An apparatus (shown in FIG. 12) may include a receiver configured to receive the codewords x₁ and x₃ (or at least the fragments of the codewords x₁ and x₃) over the wired or wireless medium. The apparatus may also include a decoder, according to an example embodiment.

FIG. 10 is a block diagram of a cross packet channel decoder 28 according to another example embodiment. In an example embodiment, the decoder 28 may be constructed to receive y₁ and y₃. y₁ may be a received vector for the first codeword x₁ or a disturbed version of x₁ (which may include noise and channel effects and may have soft values rather than binary values), and y₃ may be a received vector for the second codeword x₃ or a disturbed version of x₃ (which may include noise and channel effects and may have soft values rather than binary values). y₃ may, for example, include the combined received information from all the retransmitted packets.

The decoder 28 may, for example, include an channel decoder 502, such as a cross-packet channel decoder. The channel decoder 502 may employ the iterative principle, such as by applying different LDPC codes of different sizes to different information packets with some bits in common. The channel decoder 502 may, for example include an iterative decoder which employs the iterative principle.

For example, a channel decoder for u₁ may calculate extrinsic information for u₁, such as a posteriori reliabilities (which may include log-likelihood ratios) L₁(u₁) or û₁, from the input y₁. A channel decoder for u₃ may, for example calculate the a posteriori reliabilities L₃(u₃) or û₁ from L₁(u₁) and y₃. The reliabilities corresponding to the first codeword u₁ may then be passed back to the first channel decoder, and the process may continue iteratively for a number of iterations, which may or may not be set in advance.

FIG. 11 is a block diagram of a cross packet channel decoder 28 according to another example embodiment. The cross packet channel decoder 28 may, for example, include an iterative decoder. The decoder 28 may, in an example in which only x₁ is transmitted, iteratively decode the first packet u₁ by passing the first received codeword x₁ or y₁, or at least a fragment of the first received codeword x₁ or y₁, through a first codeword input (shown by the arrow under y₁), and pass equally probable values through a second codeword input (shown by the arrow under y₃). In an example in which both x₁ and x₃ were transmitted, the decoder 28 may iteratively decode the first packet u₁ and the second packet u₂ by passing the first received codeword x₁ or y₁, or at least the fragment of the first received codeword x₁ or y₁, through the first codeword input, and passing the second received codeword x₃ or y₃, or at least a fragment of the first received codeword x₃ or y₃, through the second codeword input.

In the example shown in FIG. 11, the decoder 28 may include a first channel decoder 504 and a second decoder 506. A first summer 508 may sum the input from the first codeword input (which may be the first codeword x₁ or y₁, or at least a fragment of the first codeword x₁ or y₁) with an output from a second summer 510. The first summer 508 may output the sum to the first decoder 504. The first decoder 504 may decode the sum received from the first summer 508, and may output the decoded sum to a first message output (shown by the arrow under L₁(u₁)) and to a third summer 512. The second summer 510 may subtract an output of the third summer 512 from a second message output received from the second decoder 506. The second summer 510 may output the difference (Δ₃) to the first summer 508 and to the third summer 512. The third summer 512 may subtract the difference (Δ₃) received from the second summer 510 from the decoded sum received from the first channel decoder 504. The third summer 512 may output the difference (Δ₁) to a fourth summer 514. The fourth summer 514 may sum the input from the second codeword input (shown by the arrow under y₃) with the difference (Δ₁) received from the third summer 512. The fourth summer 514 may output the sum to the second channel decoder 506. The second channel decoder 506 may decode the sum received from the fourth summer 514. The second channel decoder 506 may output the decoded sum to a second message output (shown by the arrow under L₃(u₃)) and to the second summer 510.

In an example embodiment, the decoder 28 may perform two sets of iterations. The decoder 28 may use “local iterations” as in an iterative algorithm such as a belief propagation decoding algorithm in each channel decoder 504, 506, which may include LDPC decoders. The decoder 28 may also use “super-iterations” to pass a posteriori reliabilities between the channel decoders 504, 506. The number of local iterations and the number of super-iterations may each be adjustable parameters for cross-packet channel decoding, according to an example embodiment.

In an example embodiment, the channel decoder 502 shown in FIG. 10 may decode the LDPC-based cross-packet channel codes with a single LDPC matrix. The decoding may be performed by, for example, constructing a codeword matrix based on the first received codeword x₁ or y₁ (or at least the fragment of the first received codeword x₁ or y₁) and the second received codeword x₃ or y₃ (or at least a fragment of the second received codeword x₃ or y₃). The decoding may also be performed by, for example, estimating the first packet u₁ and the second packet u₂ based on predefined criteria, such as minimizing an error associated with the codeword matrix. The first packet u₁ and the second packet u₂ may be determined by calculating all possible values and choosing the most probable values, or may be estimated based on algorithms such as a belief propagation decoder. Or, algorithms may estimate a plurality of possible values and the values with the lowest error may be selected, according to various example embodiments.

In an example embodiment, two constituent LDPC matrices may be combined to form a larger LDPC matrix, which may be considered the parity-check matrix, such as by amalgamation. Two separate codewords x₁ and x₂, which may be generated using separate codes, may be associated with parity check matrices H₁ and H₂. In this example,

H₁x₁=0, H₂x₂

The two codewords x₁ and x₂ may, for example, share some parity bits. These common or shared bits may be denoted x₀. Without loss of generality, these common bits may be placed at the start of both codes:

$x_{1} = {{\begin{bmatrix} x_{0} \\ x_{12} \end{bmatrix}\mspace{14mu} {and}\mspace{14mu} x_{2}} = \begin{bmatrix} x_{0} \\ x_{22\;} \end{bmatrix}}$

The parity check matrices may be partitioned into H₁=[H₁₁ H₁₂] and H₂=[H₂₁H₂₂] such that the number of columns in both H₁₁ and H₂₁ equals the length of x₀. The decoder may therefore have the following matrix stored before receiving the first codeword x₁ or y₁ (or at least the fragment of the first codeword x₁ or y₁) and the second codeword x₃ or y₃ (or at least a fragment of the first codeword x₃ or y₃):

$\quad\begin{bmatrix} H_{11} & H_{12} & 0 \\ H_{21} & 0 & H_{22} \end{bmatrix}$

If the transmission were error-free, then the following relationship would hold:

${\begin{bmatrix} H_{11} & H_{12} & 0 \\ H_{21} & 0 & H_{22} \end{bmatrix}\begin{bmatrix} x_{0} \\ x_{12} \\ x_{22} \end{bmatrix}} = 0.$

However, errors are likely to occur; the values of x₁ and x₂ may be determined by calculating all possible values and choosing the most probable values, or may be estimated based on algorithms such as a belief propagation decoder. In an example embodiment, values for x₀, x₁₂, and x₂₂ may be estimated based on the received values of y₁ and y₃. The values of x₁ and x₂ may then be determined, from which the values of u₁ and u₂ may be determined.

FIG. 12 is a block diagram of an apparatus 600 according to an example embodiment. In an example embodiment, the apparatus 600 may include a receiver 602, a decoder 604, and a memory 606. The receiver 602 may be configured to receive data, such as codewords or fragments of codewords, via a wired and/or wireless medium. In some embodiments, the apparatus 600 may also include a transmitter (not shown), which may be configured to transmit data (which may be encoded according to any of the techniques described above). In some embodiments, the receiver 602 may include a transceiver configured to both receive and transmit packets.

The apparatus 600 may also include the memory 604 configured to store codewords and/or fragments of codewords. The memory 606 may also store the parity check matrix and/or codeword matrices, according to an example embodiment.

The apparatus 600 may also include the decoder 604 configured to decode the codewords according to any of the techniques described in this disclosure. According to an example embodiment, the apparatus 600 may also include an encoder configured to encode data into codewords according to any of the techniques described in this disclosure.

FIG. 13 is a flowchart showing a method 1300 according to an example embodiment. According to an example embodiment, the method 1300 may include receiving at least a fragment of a first codeword (1302) and receiving at least a fragment of a second codeword (1304). In an example embodiments, fragments of each codeword may be put together to reconstruct the full codeword, or punctured bits may be re-inserted into the codeword. The method 1300 may further include iteratively decoding a first packet and a second packet by passing the first codeword through a first codeword input and passing the second codeword through a second codeword input (1306). In an example embodiment, the method 1300 may include decoding the first packet u₁ and the second packet u₂ by passing received codewords y₁ and y₃ through decoder 28 shown in FIG. 11.

FIG. 14 is a flowchart showing a method 1400 according to another example embodiment. According to an example embodiment, the method 1400 may include receiving at least a fragment of a first codeword (1402) and receiving at least a fragment of a second codeword (1404). The method 1400 may further include constructing a combined codeword matrix based on the first codeword matrix and the second codeword (1406). The method 1400 may further include estimating a first packet and a second packet based on predefined criteria such as minimizing an error associated with the codeword matrix (1408).

A number of embodiments of the invention have been described. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the invention. For example, the system can be extended to cross-packet coding more than two packets. Accordingly, other embodiments are within the scope of the following claims. 

1. An apparatus comprising: a receiver configured to: receive at least a fragment of a first codeword, the first codeword being based on a first packet; and receive at least a fragment of a second codeword, the second codeword being based on cross-coding the first packet and the second packet; a memory configured to store the at least the fragment of the first codeword and the at least the fragment of the second codeword; an first iterative decoder, the iterative decoder including a first codeword input and a second codeword input, the first iterative decoder being configured to iteratively decode the first packet by passing the at least the fragment of the first codeword through the first codeword input and passing equally probable values through the second codeword input; and a second iterative decoder configured to iteratively decode the second packet jointly with the first packet by using the decoded first codeword to decode the second codeword.
 2. The apparatus of claim 1, wherein the second iterative decoder is configured to iteratively decode the second packet jointly with the first packet by using the decoded first codeword as a priori information to decode the second codeword.
 3. The apparatus of claim 1, wherein the first iterative decoder and the second iterative decoder are configured to iteratively pass information associated with the first codeword to each other to decode the first codeword and the second codeword.
 4. The apparatus of claim 1, wherein the first codeword and the second codeword are coded by a linear channel code.
 5. The apparatus of claim 1, wherein the first codeword and the second codeword are coded by a low-density parity check (LDPC) code.
 6. The apparatus of claim 1, wherein the receiver is configured to receive the at least the fragment of the first codeword and the at least the fragment of the second codeword via a wireless medium.
 7. The apparatus of claim 1, further comprising a transmitter configured to transmit a jointly coded packet.
 8. The apparatus of claim 1, wherein the apparatus further includes: a first summer configured to sum the first at least the fragment with an output of a second summer and output the sum to the first iterative decoder; the second summer configured to subtract an output of a third summer from a second message output received from the second iterative decoder and output a difference to the first summer and to the third summer; the third summer configured to subtract the difference received from the second summer from the decoded sum received from the first iterative decoder and output a difference to the second summer and to a fourth summer; the fourth summer configured to sum the second at least the fragment with the difference received from the third summer and output the sum to the second iterative decoder; wherein the first iterative decoder is configured to decode the sum received from the first summer and output the decoded sum to a first message output and to the third summer; and wherein the second iterative decoder is configured to decode the sum received from the fourth summer and output the decoded difference to a second message output and to the second summer.
 9. The apparatus of claim 8, wherein: the first decoder is configured to decode the sum received from the first summer by calculating a posteriori reliabilities from the first at least the fragment; and the second channel decoder is configured to decode the difference by calculating a posteriori reliabilities from the difference received from the fourth summer and from the a posteriori reliabilities calculated by the first channel decoder.
 10. The apparatus of claim 8, wherein: the first decoder is configured to decode the sum received from the first summer by a first belief propagation decoding algorithm; and the second channel decoder is configured to decode the difference by a second belief propagation decoding algorithm.
 11. The apparatus of claim 1, further comprising a first value restorer configured to restore punctured values to the at least the fragment of the first codeword and a second value restorer configured to restore punctured values to the at least the fragment of the second codeword.
 12. An apparatus comprising: a receiver configured to: receive at least a fragment of a first codeword, the first codeword being based on a first packet; and receive at least a fragment of a second codeword, the second codeword being based on cross-coding the first packet and the second packet; a memory configured to store the at least the fragment of the first codeword and the at least the fragment of the second codeword; and a decoder configured to: construct a combined codeword matrix based on a first codeword matrix and the second codeword; and estimate the first packet and the second packet based on predefined criteria associated with the codeword matrix, the error associated with the combined codeword matrix.
 13. The apparatus of claim 12, wherein the receiver is configured to receive the at least the fragment of the first codeword and the at least the fragment of the second codeword via a wireless medium.
 14. The apparatus of claim 12, wherein the codewords are constructed from a low density parity check (LDPC) matrix.
 15. The apparatus of claim 12, wherein the parity-check matrix is based on a first codeword parity check matrix which is based on the first codeword and a second codeword parity check matrix which is based on the second codeword.
 16. The apparatus of claim 15, wherein the parity-check matrix is $\quad\begin{bmatrix} H_{11} & H_{12} & 0 \\ H_{21} & 0 & H_{22} \end{bmatrix}$ wherein H₁=[H₁₁ H₁₂], H₂=[H₂₁H₂₂], H₁x₁=0, H₂x₂=0, x₁=[x₀; x₁₂] is the first codeword, and x₂=[x₀; x₂₂] is the second codeword wherein x₀ contains common bits between x₁ and x₂.
 17. The apparatus of claim 12, wherein the decoder is further configured to construct the combined codeword based on only the at least the fragment of the first codeword and by substituting equally probable values for the at least the fragment of the second codeword during the first decoding.
 18. The apparatus of claim 12, further comprising a first value restorer configured to restore punctured values to the at least the fragment of the first codeword and a second value restorer configured to restore punctured values to the at least the fragment of the second codeword. 